In recent years, since switching elements which can withstand comparatively high current and voltage of a high frequency have been developed, most of power supply circuits which rectify commercial power supply to obtain a desired DC voltage are formed as power supply circuits of the switching type.
A switching power supply circuit reduces the size of transformers and other devices by using a high switching frequency and is used as a high-power DC-DC converter and as a power supply for various kinds of electronic apparatus.
Incidentally, generally if commercial power supply is rectified, then since current flowing through a smoothing circuit is distorted in waveform, there is a problem that the power factor indicative of the utilization efficiency of power supply is damaged.
Further, a countermeasure for suppressing harmonics generated by current of a distorted waveform is required.
Thus, a method which incorporates a so-called active filter wherein a step-up type converter of the PWM control type is provided in a rectification circuit system to make the power factor approach 1 (refer to, for example, Japanese Patent Laid-Open No. Hei 6-327246 (FIG. 11)).
A circuit diagram of FIG. 8 shows a basic configuration of such an active filter as described above.
Referring to FIG. 8, a bridge rectification circuit Di is connected to a commercial AC power supply AC. An output capacitor Cout is connected in parallel to positive/negative lines of the bridge rectification circuit Di. A rectification output of the bridge rectification circuit Di is supplied to the output capacitor Cout, and consequently, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. The DC voltage Vout is supplied as an input voltage to a load 10 such as, for example, a DC-DC converter at the following stage.
As a configuration for the power factor improvement, an inductor L, a diode D of the high speed recovery type, a resistor Ri, a switching element Q and a multiplier 11 are provided as shown in FIG. 8.
The inductor L and the diode D are inserted in series between the positive output terminal of the bridge rectification circuit Di and the positive electrode terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectification circuit Di and the negative terminal of the output capacitor Cout.
Further, in this instance, a MOS-FET is selectively used for the switching element Q, and the switching element Q is inserted between a node between the inductor L and the diode D and the primary side ground as seen in FIG. 8.
A current detection line LI and an waveform input line Lw are connected as a feedforward circuit to the multiplier 11, and a voltage detection line LV is connected as a feedback circuit to the multiplier 11.
The multiplier 11 detects the level of rectification current flowing to the negative output terminal of the bridge rectification circuit Di which is inputted from the current detection line LI.
Further, the multiplier 11 detects a rectification voltage waveform at the positive output terminal of the bridge rectification circuit Di inputted from the waveform input line Lw. This corresponds to the fact that a waveform of the commercial AC power supply AC (AC input voltage) is detected as an absolute value.
Further, the multiplier 11 detects a variation difference of the DC voltage Vout of the output capacitor Cout inputted from the voltage detection line LV. In other words, the multiplier 11 detects a variation difference of the DC input voltage to be inputted to the load 10.
Then, a drive signal for driving the switching element Q is outputted from the multiplier 11.
Rectification current which flows to the negative output terminal of the bridge rectification circuit Di is inputted from the current detection line LI to the multiplier 11. The multiplier 11 detects the rectification current level inputted from the current detection line LI. Further, the multiplier 11 detects a variation difference of the DC voltage Vout (DC input voltage) of the output capacitor Cout inputted from the voltage detection line LV. Furthermore, the multiplier 11 detects the rectification voltage waveform at the positive output terminal of the bridge rectification circuit Di inputted from the waveform input line Lw.
This corresponds to the fact that the waveform of the commercial AC power supply AC (AC input voltage) is detected as an absolute value.
The multiplier 11 first multiplies the rectification current level detected from the current detection line LI in such a manner as described above by the variation difference of the DC input voltage detected from the voltage detection line LV. Then the multiplier 11 produces a current instruction value of a waveform same as that of an AC input voltage VAC from a result of the multiplication and the waveform of the AC input voltage detected from the waveform input line Lw.
Furthermore, the multiplier 11 compares the current instruction value described above and an actual AC input current level (detected based on the input from the current detection line LI) with each other and performs PWM control with regard to the PWM signal in response to the difference to produce a drive signal based on the PWM signal. The switching element Q is switching driven with the drive signal. As a result, the AC input current is controlled so as to have a waveform same as that of the AC input voltage, and the power factor is improved so that it approaches almost 1. Further, in this instance, since the current instruction value produced by the multiplier 11 is controlled so that the amplitude thereof may vary in response to the variation difference of the DC input voltage (Vout), also the variation of the DC input voltage (Vout) is suppressed.
FIG. 9(a) illustrates the input voltage Vin and the input current Iin inputted to the active filter circuit shown in FIG. 8. The input voltage Vin corresponds to a voltage waveform as a rectification output of the bridge rectification circuit Di, and the input current Iin corresponds to a current waveform as a rectification output of the bridge rectification circuit Di. Here, although the waveform of the input current Iin has a conduction angle equal to that of the rectification output voltage (voltage Vin) of the bridge rectification circuit Di, this indicates that also the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectification circuit Di has a conduction angle equal to that of the current Iin. In other words, a power factor proximate to 1 is obtained.
FIG. 9(b) illustrates a variation of energy (power) Pchg which is inputted to and outputted from the output capacitor Cout. The output capacitor Cout accumulates energy when the input voltage Vin is high but emits energy when the input voltage Vin is low thereby to keep the flow of the output voltage.
FIG. 9(c) illustrates a waveform of charge/discharge current Ichg to/from the output capacitor Cout. The charge/discharge current Ichg is current which flows corresponding to accumulation/emission operation of the energy Pchg into/from the output capacitor Cout as can be recognized from the fact that the charge/discharge current Ichg has a phase same as that of the waveform of the input/output energy Pchg of FIG. 9(b).
Different from the input voltage Vin, the charge/discharge current Ichg has a waveform substantially same as that of a second order harmonic wave of the AC line voltage (commercial AC power supply AC). A ripple voltage Vdc is generated on the second order harmonic wave component of the AC line voltage as seen in FIG. 9(d) by a flow of energy to and from the output capacitor Cout. The ripple voltage Vdc has a phase difference of 90° from the charge/discharge current Ichg illustrated in FIG. 9(c) in order to conserve invalid energy. The rating of the output capacitor Cout is determined taking it into consideration that ripple current of the second order harmonic wave and high frequency ripple current from a boost converter switch for modulating the ripple current are processed.
FIG. 10 shows an example of a configuration of an active filter which includes the circuit configuration of FIG. 8 as a basic configuration and further includes a basic control circuit system. It is to be noted that like elements to those in FIG. 8 are denoted by like reference characters and description thereof is omitted herein.
A switching pre-regulator 17 is provided between the positive output terminal of a bridge rectification circuit Di and the positive terminal of an output capacitor Cout. The switching pre-regulator 17 is a block formed from the switching element Q, inductor L, diode D and so forth in FIG. 8.
Further, the control circuit system including a multiplier 11 further includes a voltage error amplifier 12, a divider 13 and a squarer 14.
The voltage error amplifier 12 divides a DC voltage Vout of the output capacitor Cout by means of a voltage dividing resistors Rvo-Rvd and inputs the divided voltage to the non-negated input of an operational amplifier 15. A reference voltage Vref is inputted to the negated input of the operational amplifier 15. The operational amplifier 15 amplifies a voltage of a level corresponding to an error of the divided DC voltage Vout from the reference voltage Vref with an amplification factor determined from a feedback resistor Rv1 and a capacitor Cv1 and outputs a resulting voltage as an error output voltage Vvea to the divider 13.
Further, a so-called feedforward voltage Vff is inputted to the squarer 14. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input-voltage Vin by means of an averaging circuit 16 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 14 squares the feedforward voltage Vff and outputs a resulting value to the divider 13.
The divider 13 divides the error output voltage Vvea from the voltage error amplifier 12 by the square value of the average input voltage outputted from the squarer 14 and outputs a signal as a result of the division to the multiplier 11.
In short, a voltage loop is formed from a system of the squarer 14, divider 13 and multiplier 11. Then, the error output voltage Vvea outputted from the voltage error amplifier 12 is divided by the square of the average input voltage (Vff) at a stage before it is multiplied by a rectification input signal Ivac by the multiplier 11. By this circuit, the gain of the voltage loop is maintained fixed without any variation as the square of the average input voltage (Vff). The feedforward voltage Vff has a function of open loop correction fed in a forward direction in the voltage loop.
To the multiplier 11, an output of the divider 13 obtained by dividing the error output voltage Vvea by means of the divider 13 and a rectification output (Iac) of the positive output terminal (rectification output line) of the bridge rectification circuit Di through a resistor Rvac are inputted. Here, the rectification output is indicated not as a voltage but as current (Iac). The multiplier 11 multiplies the inputs to produce and output a current programming signal (multiplier output signal) Imo. This corresponds to the current instruction value described hereinabove with reference to FIG. 8. The output voltage Vout is controlled by varying the average amplitude of the current programming signal. In particular, a PWM signal is produced in accordance with a variation of the average amplitude of the current programming signal, and switching driving is performed with a drive signal based on the PWM signal to control the level of the output voltage Vout.
Accordingly, the current programming signal has a waveform of an average amplitude for controlling the input voltage and the output voltage. It is to be noted that the active filter controls not only the output voltage Vout but also the input voltage Vin. Then, since it can be said that the current loop in the feedforward circuit is programmed with the rectification line voltage, the input to the converter (load 10) at the next stage becomes a resistive input.
FIG. 11 shows an example of a configuration of a power supply circuit wherein a current resonance type converter is connected as a next stage to an active filter having the configuration shown in FIG. 10. The power supply circuit shown in FIG. 11 is ready for the AC input voltage VAC=85 V to 288 V. In short, the power supply circuit is of the so-called wide range ready type (worldwide specifications) ready for both AC input voltages of the AC 100 V system and the AC 200 V system as commercial AC power supply. Further, the load power for which the power supply circuit is ready is 600 W or more. Further, the current resonance type converter adopts a configuration by the separately excited half bridge coupling system.
The power supply circuit shown in FIG. 11 is provided in display apparatus such as television receivers and monitor apparatus which include a plasma display panel which has been and is being popularized in recent years. In other words, the power supply circuit shown in FIG. 11 supplies operating power supply for an internal circuit of a display apparatus (plasma display apparatus) which includes such a plasma display panel as described above.
To a commercial AC power supply AC line in this instance, two common mode choke coils CMC, CMC and three across capacitors CL are connected in a connection scheme shown in FIG. 11 to form a line noise filter for common mode noise.
Further, in this instance, a main switch SW for activating/deactivating the power supply is shown inserted in the commercial AC power supply AC line.
To the positive/negative lines of the commercial AC power supply AC, the positive input terminals and the negative input terminals of two bridge rectification circuits Di1 and Di2 are connected commonly, respectively. Further, the positive output terminals of the bridge rectification circuits Di1 and Di2 are connected to each other, and the negative output terminals (ground terminals) of the bridge rectification circuits Di1 and Di2 are connected to each other. In short, in this instance, two stages of bridge rectification circuits are provided for the commercial AC power supply AC.
Further, a normal mode noise filter 4 formed from one choke coil LN and three filter capacitors (film capacitors) CN, CN, CN connected in such a manner as shown in FIG. 11 is connected between the positive output terminals and the negative output terminals (primary side ground) of the bridge rectification circuits Di1 and Di2.
An active filter circuit 8 is provided at a next stage to the normal mode noise filter 4.
The active filter circuit 8 is based on the configuration described hereinabove with reference to FIG. 10. In particular, the active filter circuit 8 includes a step-up type converter of the PWM control type which performs switching between the rectification outputs inputted from the bridge rectification circuits Di1 and Di2. Such a step-up type converter as mentioned above is formed, for example, including a switching element, and a control circuit system for driving the switching element in accordance with a PWM control system.
Further, in order to cope with a heavy load condition of, for example, the load power Po=600 W or more as in the present case, such a countermeasure that a plurality of switching elements are provided and connected in parallel or the like is adopted. When the load is heavy, particularly in a condition that the AC input voltage VAC is 100 V or less, very high current flows through a switching element. Therefore, a plurality of switching elements are connected in parallel in such a manner as described above so that the peak level of switching current flow to each of the switching devices is suppressed. Consequently, the reliability of the active filter circuit 8 is enhanced.
Meanwhile, the control circuit system includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, a drive circuit for outputting a drive signal for switching driving the switching elements, and so forth, and is formed, for example, as a single IC chip. A circuit block corresponding to the multiplier 11, voltage error amplifier 12, divider 13, squarer 14 and so forth shown in FIG. 10 is incorporated in the IC as the control circuit system. And, the feedback circuit system and the feedforward circuit system are connected in such a manner as described hereinabove with reference to FIGS. 8 and 10 to the IC chip as the control circuit system, and the IC chip as the control circuit system drives the switching elements by PWM control based on feedback outputs from the circuit systems.
The switching driving of the switching elements in the active filter circuit 8 having the configuration described above is performed in accordance with the drive signal based on the PWM control so that the conduction angle of the rectification output current may be substantially equal to that of the rectification output voltage waveform as described hereinabove with reference to FIGS. 8 and 10. That the conduction angle of the rectification output current is substantially equal to that of the rectification output voltage waveform signifies that the conduction angle of the AC input current flowing in from the commercial AC power supply AC is substantially equal to that of the waveform of the AC input voltage VAC, and as a result, the power factor is controlled so as to approach 1. In short, improvement of the power factor is achieved. In an actual case, a characteristic that a power factor PF=approximately 0.995 is obtained when the load power Po=600 W.
Further, the active filter control circuit 8 shown in FIG. 11 operates also such that the average value of the rectification smoothed voltage Ei (corresponding to Vout in FIG. 10) may be a fixed voltage within the range of AC input voltage VAC=85 V to 288 V. In short, a DC input voltage stabilized to 375 V is supplied to the current resonance type converter at the next stage irrespective of the range of variation of the AC input voltage VAC=85 V to 264 V.
The range of the AC input voltage VAC=85 V to 288 V continuously covers the AC 100 V system and the AC 200 V system of commercial AC power supply. Accordingly, the stabilized DC input voltage (Ei) of the equal level is supplied to the switching converter at the next stage irrespective of whether the commercial AC power supply AC is of the 100 V system or the 200 V system. In short, the power supply circuit shown in FIG. 11 is formed also as a power supply circuit ready for a wide range through the provision of the active filter.
In this instance, a set of three smoothing capacitors CiA, CiB and CiC are connected in series at the next stage to the active filter circuit 8.
The set of the smoothing capacitors [CiA//CiB//CiC] corresponds to the output capacitor Cout in FIGS. 8 and 10. Accordingly, in this instance, the rectification smoothed voltage Ei is obtained as a voltage across the set of the smoothing capacitors [CiA//CiB//CiC] connected in parallel. The rectification smoothed voltage Ei is supplied as a DC input voltage to converter sections 201, 202 and 203 at the next stage. Then, as described hereinabove, the voltage (rectification smoothed voltage Ei) across the smoothing capacitors [CiA//CiB//CiC] in this instance is stabilized at 375 V.
Further, in the power supply circuit shown in FIG. 11, in order to cope with such a heavy load condition as described hereinabove, a plurality of composite resonance type converts which use the DC input voltage of the rectification smoothed voltage Ei as the operating voltage are provided. The composite resonance type converter here signifies a switching converter of a configuration wherein, in addition to a resonance circuit for making operation of the switching converter that of a resonance type, another resonance circuit is added to the primary side or the secondary side such that the plurality of resonance circuits operate compositely in one switching converter. In FIG. 11, three composite resonance type converters of the first converter section 201, second converter section 202 and third converter section 203 are provided. Each of the composite resonance type converters here is formed from a primary side partial voltage resonance circuit added to a current resonance type converter as hereinafter described.
For example, the first converter section 201 includes, as components thereof, two switching elements Q1 and Q2 as shown in FIG. 11. In this instance, the switching elements Q1 and Q2 are connected in a half-bridge connection such that the switching element Q1 serves as a high side switching element and the switching element Q2 serves as a low side switching element, and are connected in parallel to the rectification smoothed voltage Ei (DC input voltage). In short, the first converter section 201 has a configuration as a current resonance type converter of the half-bridge coupling type.
The current resonance type converter in this instance is of the separately excited type, and corresponding to this, a MOS-FET is used for the switching elements Q1 and Q2. Clamp diodes DD1 and DD2 are connected in parallel to the switching elements Q1 and Q2, respectively, such that a switching circuit is formed. The clamp diodes DD1 and DD2 form paths along which reverse current upon turning off of the switching elements Q1 and Q2 flows, respectively.
A control IC 2 includes an oscillation circuit for driving the current resonance type converter in a separately excited manner, a control circuit, a protection circuit and the like. The control IC 2 is an analog IC (Integrated Circuit) having a bipolar transistor inside thereof.
The control IC 2 operates with a DC voltage inputted to a power supply input terminal Vcc. In this instance, the rectification smoothed voltage Ei inputted through a resistor Rs is inputted to the power supply input terminal Vcc. Further, a ground electrode E is directly connected to the primary side ground.
The control IC 2 includes two drive signal output terminals VGH and VGL as terminals for outputting a drive signal (gate voltage) to the switching elements.
A drive signal for switching driving the high side switching element is outputted from the drive signal output terminal VGH, and another drive signal for switching driving the low side switching element is outputted from the drive signal output terminal VGL.
In this instance, the drive signal output terminal VGH is connected to the gate of the high side switching element Q1. Meanwhile, the drive signal output terminal VGL is connected to the gate of the low side switching element Q2.
Consequently, the drive signal for the high side outputted from the drive signal output terminal VGH is applied to the gate of the switching element Q1, and the drive signal for the low side outputted from the drive signal output terminal VGL is applied to the gate of the switching element Q2.
The control IC 2 produces an oscillation signal of a required frequency from an internal oscillation circuit. Then, the control IC 2 utilizes the oscillation signal produced by the oscillation circuit to produce a drive signal for the high side and another drive signal for the low side. Here, the drive signal for the high side and the drive signal for the low side are produced in such a mutual relationship that they have a phase difference of 180°. Then, the drive signal for the high side is outputted from the drive signal output terminal VGH, and the drive signal for the low side is outputted from the drive signal output terminal VGL.
Since the drive signal for the high side and the drive signal for the low side are applied to the switching elements Q1 and Q2, respectively, within a period within which the drive signal has the H level, the gate voltage of the switching element Q1 or Q2 becomes equal to or higher than a gate threshold value and the switching element Q1 or Q2 is placed into an on state. On the other hand, within another period within which the drive signal has the L level, the gate voltage becomes equal to or lower than the gate threshold value and the switching element Q1 or Q2 is placed into an off state. Consequently, the switching elements Q1 and Q2 are switching driven with a required switching frequency at timings at which they are turned on/off alternately.
To an activation terminal Vt of the control IC 2, a starting signal Vt1 outputted from a microcomputer (not shown in FIG. 11 (not shown) provided in an apparatus in which the power supply circuit shown in FIG. 11 is incorporated is inputted.
The control IC 2 is activated at a timing at which the starting signal is inputted to start operation thereof. In short, the control IC 2 starts outputting of the drive signals from the drive signal output terminal VGH and the drive signal output terminal VGL. Accordingly, the operation starting timing of the first converter section 201 is determined by an inputting timing of the starting signal Vt1 of the control IC 2.
An insulating converter transformer PIT-1 is provided for transmitting the switching outputs of the switching elements Q1 and Q2 from the primary side to the secondary side.
A primary winding N1 of the insulating converter transformer PIT-1 is connected at an end portion thereof to the node (switching output point) of the switching elements Q1 and Q2 through a primary side series resonance capacitor C1 and at the other end portion thereof to the primary side ground. Here, a primary side series resonance circuit is formed from the capacitance of the primary side series resonance capacitor C1 and the leakage inductance (L1) of the primary winding N1. The primary side series resonance circuit performs resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied thereto, and thereby makes operation of the switching circuit formed from the switching elements Q1 and Q2 that of the current resonance type.
A partial resonance capacitor Cp is connected in parallel between the drain-source of the switching element Q2. The capacitance of the partial resonance capacitor Cp and the current detection line LI of the primary winding N1 cooperatively form a parallel resonance circuit (partial voltage resonance circuit). Then, partial voltage resonance operation wherein voltage resonance occurs only upon turning on of the switching elements Q1 and Q2 is obtained.
In this manner, the power supply circuit shown in FIG. 11 has a form as a composite resonance type converter wherein a resonance circuit for making a primary side switching converter that of the resonance type is combined with another resonance circuit.
On the secondary side of the insulating converter transformer PIT-1, two secondary windings N2a and N2b are wound independently of each other as secondary windings.
The secondary winding N2a in this instance has a center tap provided thereon as shown in FIG. 11 and connected to the secondary side ground, and a full-wave rectification circuit formed from rectification diodes Do1 and Do2 and a smoothing capacitor Co1 is connected to the secondary winding N2a. Consequently, a secondary side DC output voltage Eo1 is obtained as a voltage across the smoothing capacitor Co1. The secondary side DC output voltage Eo1 is supplied to the load side not shown and is branched and inputted also as a detection voltage for a control circuit 1.
The control circuit 1 supplies a voltage or current whose level is adjusted in response to the level of the secondary side DC output voltage Eo1 inputted thereto as a control output to a control input terminal Vc of the control IC 2. The control IC 2 adjusts, for example, the frequency of the oscillation signal in response to the control output inputted to the control input terminal Vc to adjust the frequency of the drive signals to be outputted from the drive signal output terminals VGH and VGL. Consequently, the switching frequency of the switching elements Q1 and Q2 is variably controlled, and as the switching frequency is adjusted in this manner, the level of the secondary side DC output voltage Eo1 is controlled so as to be fixed. In other words, stabilization according to the switching frequency control method is performed.
Further, in this instance, the circuit is formed such that the secondary side DC output voltage Eo1 is branched to form secondary side output voltages Eo and Eo2.
The circuit system for producing the secondary side output voltage Eo is formed as a step-down type converter wherein a switching element Q7 formed from a MOS-FET, a rectification diode Dcn1, a choke coil L1 for high frequency noise removal, a smoothing capacitor Co, and a control circuit 7 for executing PWM (Pulse Width Modulation) control are connected in such a manner as seen in FIG. 11.
The switching element Q7 is switching driven by the control circuit 7 to switch the secondary side DC output voltage Eo1 to obtain an alternating output. The alternating output is rectified and smoothed by a half-wave rectification circuit formed from the choke coil L1, rectification diode Dcn1 and smoothing capacitor Co to produce the secondary side DC output voltage Eo as a voltage across the smoothing capacitor Co.
Here, the control circuit 7 executes the PWM control in response to the level of the secondary side DC output voltage Eo. Consequently, the switching operation of the switching element Q7 is controlled so that the switching frequency is fixed in response to the level of the secondary side DC output voltage Eo and the on-period within one switching period is varied. Consequently, the level of the secondary side DC output voltage Eo is controlled so as to be fixed. In other words, stabilization of the secondary side DC output voltage Eo is achieved.
Also the circuit system for producing the secondary side output voltage Eo2 is formed as a step-down type converter wherein a switching element Q8 formed from a MOS-FET, a rectification diode Dcn2, a choke coil L2, a smoothing capacitor Co2 and a control circuit 7 are connected in a similar connection scheme as that of the circuit system for producing the secondary side DC output voltage Eo1 described hereinabove.
Accordingly, also in this instance, a secondary side DC output voltage Eo2 stabilized by the PWM control of the control circuit 7 is obtained as a voltage across the smoothing capacitor Co2.
Meanwhile, for the secondary winding N2b, a full-wave rectification circuit formed from a bridge rectification circuit DBR and a smoothing capacitor Co3 is formed, and a secondary side DC output voltage Eo3 is obtained as a voltage across the smoothing capacitor Co3 by rectification smoothing of the full-wave rectification circuit.
The second converter section 202 has a configuration as a composite resonance type converter wherein a current resonance type converter and a primary side partial voltage resonance circuit are combined by connecting switching elements Q3 and Q4 connected in a half-bridge connection, clamp diodes DD3 and DD4, a partial resonance capacitor Cp, a control IC 2, a primary winding N1 of an insulating converter transformer PIT-2 and so forth are connected in a connection scheme similar to that of the first converter section 201 described hereinabove.
Further, the secondary side of the second converter section 202 is connected at a center tap of the secondary winding N2 to the secondary side ground, and a full-wave rectification circuit including rectification diodes Do1 and Do2, smoothing capacitors Co4 and Co5 and a resistor R1 for noise removal is formed in such a manner as seen in FIG. 11 for the secondary winding N2. Consequently, a secondary side DC output voltage Eo4 is produced as a voltage across the smoothing capacitor Co5.
Further, in the second converter section 202, since the control circuit 7 executes switching frequency control of the primary side converter based on the level of the secondary side rectification smoothed voltage obtained across the smoothing capacitor Co4, stabilization of the secondary side DC output voltage Eo4 is achieved.
Further, in the second converter section 202, an activation signal Vt3 outputted from the microcomputer is inputted to an activation terminal Vt of the control IC 2.
Also the third converter section 203 has a configuration as a composite resonance type converter wherein a current resonance type converter and a primary side partial voltage resonance circuit are combined by connecting switching elements Q5 and Q6 connected in a half-bridge connection, clamp diodes DD5 and DD6, a partial resonance capacitor Cp, a control IC 2, an insulating converter transformer PIT-3 (primary winding N1 and secondary winding N2, rectification diodes Do1 and Do2, smoothing capacitors Co6 and Co7 and a resistor R2 in a connection scheme similar to that of the second converter section 202. Also in the third converter section 203, a secondary side DC output voltage Eo5 stabilized by switching frequency control by the control circuit 7 is obtained.
Further, an activation signal Vt2 outputted from the microcomputer is inputted to an activation terminal Vt of the control IC 2 of the third converter section 203.
In the configuration described above, the six secondary side DC output voltages Eo, Eo1 to Eo5 are obtained from the secondary side. The secondary side DC output voltages have, for example, such applications and load specifications as given below.                Eo: logic power supply, 5 V/6 A to 2 A        Eo1: analog IC driving power supply, 12 V/0.4 A        Eo2: digital IC driving power supply, 3.3 V/1.5 A        Eo3: sound outputting power supply, 26 V/1.3 A to 0.1 A        Eo4: data power supply, 70 V/2.5 A to 0.35 A        Eo5: maintaining power supply, 200 V/1.75 A to 0.1 A        
Further, the maximum load powers for which the individual converter sections should be ready are,
first converter section 201: 75 W,
second converter section 202: 175 W and
third converter section 203: 350 W and totally 600 W.
Further, the cores of the insulating converter transformers are selected in the following manner in accordance with such maximum load powers for which the converter sections should be ready as mentioned hereinabove:
PIT-1: EER-35
PIT-2: EER-40
PIT-3: EER-42
Meanwhile, the choke coils L11 and L12 of the step-down type converters employ a ferrite core of EE-25.
The power supply circuit provided in the plasma display apparatus outputs a plurality of secondary side DC output voltages Eo, Eo1 to Eo5 individually corresponding to the different load conditions as illustrated in FIG. 11. Further, particularly in the plasma display apparatus, for the convenience of the circuit configuration, when the power supply is activated to start the DC input voltage (rectification smoothed voltage Ei (375 V)), it is necessary for the secondary side DC output voltages to be started in a predetermined order.
More particularly, the secondary side DC output voltage Eo which is the logic power supply is started up first, and then the secondary side DC output voltage Eo5 which is the maintaining power supply and the secondary side DC output voltage Eo4 which is the data power supply are successively started.
Thus, in order to obtain such a starting order of the secondary side DC output voltages as described above, the microcomputer outputs the starting voltages Vt1, Vt2 and Vt3 to the starting terminals Vt of the control ICs 2 of the converter sections (201, 202 and 203). Control operation of the starting up order of the secondary side DC output voltages by the starting voltages Vt1, Vt2 and Vt3 is illustrated in a timing chart of FIG. 12.
Here, the power supply circuit shown in FIG. 11 has a configuration of a so-called main power supply, and no standby power supply is shown in FIG. 11. Since the microcomputer is supplied with the standby power supply, even if the main power supply is not in an activated state, the microcomputer can operate.
Then, if the main switch SW is changed over from an off state to an on state in order to activate the circuit shown in FIG. 11 which is a main power supply, then the commercial AC power supply AC is supplied to the circuit and the rectification smoothed voltage Ei is obtained. Then, if it is detected by the microcomputer that the rectification smoothed voltage Ei rises up to a prescribed level (for example, 375 V), then the microcomputer changes over the starting signal Vt1 from the L level to the H level and outputs the starting signal Vt1 of the H level at a timing of time t1. Consequently, the control IC 2 of the first converter section 201 to which the starting signal Vt1 is inputted starts switching driving operation at time t1. Then, in response to this, the secondary side DC output voltage Eo obtained on the secondary side of the first converter section 201 starts its rise from the 0 level at time t1 and rises up to a prescribed level (5 V) at a point of time when a certain period of time elapses. Thereafter, the secondary side DC output voltage Eo maintains the stabilized state at 12 V by the constant voltage control operation by the step-down converter.
It is to be noted that it is described for the confirmation that also the remaining secondary side DC output voltages Eo1, Eo2 and Eo3 produced by the same first converter section 201 rise at a timing substantially same as that of the secondary side DC output voltage Eo.
Then, the activation signal Vt2 is set such that, at time t2 after the secondary side DC output voltage Eo rises up-to and becomes stable at the prescribed level after the rise starts at time t1 as described above, it is changed over from the L level to the H level and outputted as a H level signal.
Consequently, the control IC of the third converter section 203 is activated at time t2. In response to this, the secondary side DC output voltage Eo5 starts its rise from the 0 level later than time t2, and at a point of time when a certain period of time elapses, the secondary side DC output voltage Eo5 is fixed at a prescribed level (200 V).
Further, at time t3 after the secondary side DC output voltage Eo5 is placed into a stabilized state at the prescribed level as described above, the microcomputer changes over the activation signal Vt3 from the L level to the H level. In response to this, the control IC of the second converter section 202 is activated at time t3, and the secondary side DC output voltage Eo4 rises such that it starts its rise from the 0 level after time t3 and is fixed at a prescribed level (70 V) at a point of time when a certain period of time elapses.
In this manner, the power supply circuit shown in FIG. 11 controls the rise time of the secondary side DC output voltage so that appropriate activation operation as a power supply circuit can be obtained.
As can be recognized from the foregoing description, the power supply circuit shown in FIG. 11 as a related art is formed incorporating an active filter having the conventionally known configuration shown in FIGS. 8 and 10 as a basic configuration. Further, in the case of the circuit shown in FIG. 11, three composite resonance type converters are connected in parallel at the next stage to the active filter. Furthermore, a step-down type converter for obtaining the secondary side DC output voltages Eo and Eo2 is provided in the composite resonance type converter of the first converter section 201.
Such a configuration as described above is adopted to achieve improvement of the power factor. Further, the power supply circuit shown in FIG. 11 is ready for a so-called wide range in such a manner that it operates with the AC 100 V system and the AC 200 V system as the commercial AC power supply. Further, the circuit system for constant voltage control by the switching control method and a required number of step-down type converters provided on the secondary side are combined to achieve stabilization of the secondary side DC output voltage.
However, the power supply circuit having the configuration shown in FIG. 11 has the following problem.
The power conversion efficiency of the power supply circuit shown in FIG. 11 is given as a synthesized value of the AC-DC power conversion efficiency (ηAC→DC) and the DC-DC power conversion efficiency (ηDC→DC) of the current resonance type converters (first, second and third converter sections 201, 202 and 203) at the succeeding stage.
Here, the DC-DC power conversion efficiency (ηDC→DC) of the first, second and third converter sections 201, 202 and 203 is approximately 95%.
Meanwhile, the AC-DC power conversion efficiency (ηAC→DC) of the active filter is 93% when the AC input voltage VAC=100 V and 95% when the AC input voltage VAC=230 V.
Accordingly, the combined power conversion efficiency is, when the AC input voltage VAC=100 V,93%×95%=88.3%and when the AC input voltage VAC=230 V,95%×95%=90.2%
Corresponding to this, the AC input power is 679.5 W when the AC input voltage VAC=100 V, and 665.2 W when the AC input power is 230.
In short, when the AC input voltage VAC=100 V (AC 200 V system), the conversion efficiency on the active filter circuit side drops and the overall efficiency drops when compared with that when the AC input voltage VAC=230 V (AC 100 V system).
Further, it is necessary to design the circuit shown in FIG. 11 such that the AC-DC power conversion efficiency (ηAC→DC) of the active filter may maintain 94% to 97%, for example, within the range of AC input voltage VAC=100 V to 230 V so that it may not become lower than the above-described characteristic of the power conversion efficiency under the condition that the load power Po=600 W or more.
Further, while, in the active filter circuit 8, switching operation as a step-up type converter is performed, since the switching operation depends upon dv/di and di/dt and is hard switching operation, the generation level of noise is very high.
Further, since three composite resonance type converters and two step-down type converts are provided at the next stage to the active filter, also switching noise of them is so high that it cannot be ignored. Particularly since the step-down type converters perform hard switching operation, the amount of switching noise generation is great. In contrast, although the composite resonance type converters perform soft switching operation and generates switching noise of a smaller amount than the hard switching converters, since the arrangement includes three composite resonance type converters, the overall noise amount is great as much. From them, a comparatively heavy noise suppression countermeasure is required.
Then, from the necessities described, the power supply circuit shown in FIG. 11 first includes two bridge rectification circuits Di1 and Di2 in the rectification circuit system for rectifying the commercial AC power supply AC.
Further, it is necessary to provide a plurality of power choke coils in the active filter circuit 8. Furthermore, with regard to a semiconductor element for switching, it is necessary to connect a plurality of switching elements (transistors, diodes or the like) in parallel and add a drive circuit so that the switching elements may be driven appropriately. Further, also it is necessary to attach a large-size heat radiating plate to the semiconductor elements.
Furthermore, in the circuit shown in FIG. 11, a line noise filter including two common mode choke coils and three across capacitors is formed for the line of the commercial AC power supply AC. In short, two or more stages of line noise filters are required.
Further, the normal mode noise filter 4 formed from a choke coil LN and three filter capacitors CN is provided for the rectification output line. Furthermore, in the active filter circuit 8, also it is necessary to provide an RC snubber circuit for the switching elements. Particularly in order to be ready for a heavy load as in the case of the circuit of FIG. 11, the resistor which forms the RC snubber is a cement resistor and has a large size.
In this manner, in an actual circuit, a countermeasure against noise based on a very large number of parts is required, and this gives right to increase of the cost and increase of the mounting area of the power supply circuit board.
Furthermore, in the circuit shown in FIG. 11, it is considered that three kinds of switching converters exist in a mixed manner. In particular, they are a step-up type converter in the active filter circuit 8, a composite resonance type converter which forms the first to third converter sections 201 to 203 and a step-down type converter added to the first converter section 201.
In this instance, while the switching frequency of the step-up converter of the active filter circuit 8 is 50 KHz, the switching frequency of the composite resonance type converters of the first to third converter sections 201 to 203 is within the range from 70 KHz to 150 KHz. Further, the step-down converter of the first converter section 201 has a switching frequency of, for example, 100 KHz.
Where the switching frequencies of the switching converters are different from each other in this manner, there is a problem also that the ground potentials of the primary side and the secondary side interfere with each other and the operation of the power supply circuit is liable to be rendered unstable.
Further, the power supply circuit shown in FIG. 11 has a configuration that it includes the three converter sections 201, 202 and 203 each in the form of a composite resonance type converter. This arises from the fact that, as described hereinabove with reference to FIG. 12, in this instance, the rising timing of the secondary side DC output voltage must be controlled at the three stages of the times t1, t2 and t3.
In short, the starting signal is outputted as a signal (activation control signal) for activating the control IC 2. Accordingly, in order to implement a starting sequence of the secondary side DC output voltage corresponding to the times t1, t2 and t3 illustrated in FIG. 12, three control ICs which are activated in response to the starting voltages Vt1, Vt2 and Vt3 are required correspondingly. Therefore, three composite resonance type converters are provided corresponding to the three control ICs 2, and secondary side DC output voltages to be produced by the three composite resonance type converters are allocated in accordance with the order of the required starting sequence.
However, with the configuration wherein the secondary side DC output voltages are successively started in such a manner as described above, a number of control ICs corresponding to the number of starting signals are required, and accordingly, a number of converter sections corresponding to the number of starting signals are required. In short, this signifies that, in response to an increase of the number of stages of a starting sequence of secondary side DC output voltages, also it is necessary to increase the number of converter sections.
This gives rise to a disadvantage that, if it is tried to cope with an increase of the number of stages of a starting sequence of secondary side DC output voltages, then the number of converter sections increases only for this, and as a result, the number of components of the converter sections beginning with a control IC, an insulating converter transformer PIT, a switching element and so forth increases exceeding a necessary number. Such increase of the number of converter sections is not preferable because it results in increase in size and weight of a power supply circuit board. Further, where the converter sections increase, also the switching loss of the primary side switching elements increases correspondingly, and this is disadvantageous also in power conversion efficiency.